Amplifier circuit and methods of operation thereof

ABSTRACT

A signal amplifying circuit and associated methods and apparatuses, the circuit comprising: a signal path extending from an input terminal to an output terminal, a gain controller arranged to control the gain applied along the signal path in response to a control signal; an output stage within the signal path for generating the output signal, the output stage having a gain that is substantially independent of its supply voltage, and a variable voltage power supply comprising a charge pump for providing positive and negative output voltages, the charge pump comprising a network of switches that is operable in a number of different states and a controller for operating the switches in a sequence of the states so as to generate positive and negative output voltages together spanning a voltage approximately equal to the input voltage.

This is a continuation of application Ser. No. 16/547,063, filed on Aug.21, 2019, now U.S. Pat. No. 11,031,863, which is a continuation ofapplication Ser. No. 15/597,900, filed on May 17, 2017, now U.S. Pat.No. 10,587,187, which is a continuation of application Ser. No.15/085,360, filed on Mar. 30, 2016, now U.S. Pat. No. 9,685,855, whichis a continuation of application Ser. No. 14/174,553, filed on Feb. 6,2014, now U.S. Pat. No. 9,306,448, which is a continuation ofapplication Ser. No. 13/152,770, filed on Jun. 3, 2011, now U.S. Pat.No. 8,660,277, which is a continuation of application Ser. No.12/390,235, filed Feb. 20, 2009, now abandoned, which is a continuationof application Ser. No. 12/000,549, filed Dec. 13, 2007, now U.S. Pat.No. 7,714,660, which claims priority to United Kingdom Application No.0625955.0, filed Dec. 22, 2006, the disclosures of which are herebyincorporated by reference in their entireties.

The present invention relates to circuitry for improving the efficiencyof an amplifier. The invention further relates to a method for improvingthe efficiency of an amplifying circuit.

When receiving information signals, such as audio signals for example,for outputting to one or more transducers, such as a speaker forexample, the information signals generally need to be adjusted inamplitude. One method of achieving this adjustment includes using acontrol signal, an example of such a signal being a gain control signal,which varies the gain, and thus the amplitude, of the information signalprior to outputting to the transducer.

FIG. 1 illustrates an example of a known amplifier 10.

The amplifier 10 comprises a gain controller 20; an output stage 40; anda power supply 60.

The gain controller 20 receives an input information signal S1 and aninput control signal S2. The control signal S2 controls the gaincontroller 20 that outputs a gain controlled information signal S3 whichis fed as an input signal into the output stage 40. The output stage 40outputs an output signal S4 that is used to drive a load 70.

The output stage 40 and the gain controller 20 are supplied by the powersupply 60 which takes power from some external power source and suppliesdual, fixed level, supply voltages +V1 and −V1.

The amplitude of the output signal S4 that drives the load 70 is variedi.e. amplified or attenuated, in response to the control input signalS2, by the combined gain of the gain controller 20 and output stage 40.

The power efficiency of the amplifier 10, i.e. the ratio of the powerdelivered to the load to the power taken from the power source, is animportant parameter of the amplifier. It impacts both power consumption,which is important in battery-powered systems for example, and powerdissipation, which influences cost in terms of heatsinking for example.

There are thus advantages in methods and circuits for improving theefficiency of amplifiers such as amplifier 10.

In a first aspect of the invention there is provided a signal amplifyingcircuit comprising:

-   -   A main input terminal for receiving an input signal;    -   A main output terminal for outputting an output signal;    -   a signal path extending from the main input terminal to the main        output terminal;    -   a gain controller arranged to control the gain applied along the        signal path in response to a control signal;    -   an output stage within the signal path for generating the output        signal, the output stage having a gain that is substantially        independent of its supply voltage,    -   a variable voltage power supply comprising a charge pump circuit        for providing a plurality of output voltages, the charge pump        circuit comprising:        -   an supply input terminal and a common terminal for            connection to an input voltage,        -   first and second supply output terminals for outputting the            plurality of output voltages, the supply output terminals in            use being connected to the common terminal via respective            first and second reservoir capacitors,        -   first and second flying capacitor terminals for connection            to a flying capacitor,        -   a network of switches that is operable in a plurality of            different states for interconnecting the terminals, and        -   a controller for operating the switches in a sequence of the            states, the sequence being adapted repeatedly to transfer            packets of charge from the supply input terminal to the            reservoir capacitors via the flying capacitor depending on            the state, and thereby generating positive and negative            output voltages together spanning a voltage approximately            equal to the input voltage, and centered on the voltage at            the common terminal.    -   wherein the variable voltage power supply is arranged to vary a        supply voltage of the output stage in response to the control        signal.

The variable voltage power supply may comprise an input selector forselecting a signal to be input into the input supply terminal of thecharge pump circuit, depending on a signal derived from the controlsignal, therefore controlling the voltage levels at the first and secondsupply output terminals of the variable voltage power supply. The switchnetwork may be operable in at least a first state and a second state,the controller being adapted to operate the switches in a sequence whichinterleaves repetitions of the first and second states, the first statebeing effective to divide the input voltage between the flying capacitorand first reservoir capacitor in series, the second state beingeffective to apply the flying capacitor's portion of the divided voltageacross the second reservoir capacitor. In the first state, the first andsecond flying capacitor terminals may be connected to the supply inputterminal and the first supply output terminal respectively, and in thesecond state the first and second flying capacitor terminals may beconnected to the common terminal and the second supply output terminalrespectively.

The switch network may be further operable in a third state effective toapply the flying capacitor's portion of the divided voltage across thefirst reservoir capacitor, and wherein the controller may be adapted toinclude repetitions of the third state within the sequence. In the thirdstate, the first and second flying capacitor terminals may be connectedto the first supply output terminal and the common terminalrespectively. The controller may be adapted to include the third stateless frequently than the first and second states.

The switch network may be operable in at least a fourth state and afifth state, the fourth state being effective to charge up the flyingcapacitor to the input voltage, the fifth state being effective todivide the voltage on the flying capacitor between the first reservoircapacitor and second reservoir capacitor in series, and wherein thecontroller may be adapted to operate the switches in a sequence whichinterleaves repetitions of the fourth and fifth states. In the fourthstate, the first and second flying capacitor terminals may be connectedto the supply input terminal and the common terminal respectively, andin the fifth state, the first and second flying capacitor terminals maybe connected to the first supply output terminal and the second supplyoutput terminal respectively.

The switch network may be operable to connect the first flying capacitorterminal independently to any of the supply input terminal, the firstsupply output terminal and the common terminal.

The switch network may be operable to connect the second flyingcapacitor terminal independently to any of the first supply outputterminal, the common terminal and the second supply output terminal.

The switch network may comprise:

-   -   a first switch for connecting the supply input terminal to the        first flying capacitor terminal,    -   a second switch for connecting the first flying capacitor        terminal to the first supply output terminal,    -   a third switch for connecting the first flying capacitor        terminal to the common terminal,    -   a fourth switch for connecting the second flying capacitor        terminal to the first supply output terminal,    -   a fifth switch for connecting the second flying capacitor        terminal to the common terminal, and    -   a sixth switch for connecting the second flying capacitor        terminal to the second supply output terminal.

The controller may be operable to control the network to generate thesplit rail supply with positive and negative output voltages togetherspanning a voltage approximately equal to the input voltage, andcentered on the voltage at the common terminal when it may be operatingin a first mode, the circuit being further operable in a second mode toyield positive and negative output voltages each up to substantially theinput voltage across the intermediate supply terminals.

The variable voltage power supply may vary the supply voltage of theoutput stage by having the charge pump circuit switch between the firstmode and the second mode in response to the control signal.

The controller may be adapted in the second mode to operate the switchesin a sequence which interleaves repetitions of at least second and sixthstates, the sixth state being effective to charge the flying capacitorand the first reservoir capacitor substantially to the input voltage,the second state being effective to transfer the voltage from the flyingcapacitor to the second reservoir capacitor. In the second state, thefirst and second flying capacitor terminals may be connected to thecommon terminal and the second supply output terminal respectively, andin the sixth state the first flying capacitor terminal may be connectedto both the supply input terminal and the first supply output terminaland the second flying capacitor terminal may be connected to the commonterminal.

The controller may be adapted in the second mode to include in thesequence repetitions a seventh state, the seventh state being effectiveto charge the flying capacitor independent of either reservoircapacitor. In the seventh state the first flying capacitor terminal maybe connected to the supply input terminal only and the second flyingcapacitor terminal may be connected to the common terminal.

The network may include a switch which may be used in the second mode toconnect the supply input terminal to the first supply output terminalindependently of the first flying capacitor terminal. The switch may bealways closed when the circuit is operating in a particularimplementation of the second mode, thus ensuring that the firstreservoir capacitor is always connected between the supply inputterminal and the common terminal when operating in this particularimplementation.

The controller may be operable to implement the second mode of operationin any of the variants herein described.

The controller may be adapted to vary the sequence of states accordingto load conditions.

The charge pump circuit may be arranged to operate in a closed loopconfiguration. The first reservoir capacitor may be charged only whenthe voltage at the first supply output terminal falls below a firstthreshold value and the second reservoir capacitor may be charged onlywhen the voltage at the second supply output terminal falls below asecond threshold value. Alternatively, the first reservoir capacitor andthe second reservoir capacitor may be both charged only when either thevoltage at the first supply output terminal falls below a firstthreshold value or the voltage at the second supply output terminalfalls below a second threshold value. The variable voltage power supplyfurther may comprise at least one comparator for comparing the voltageat each of the supply output terminals with at least one referencevoltage.

The at least one reference voltage may depend on a signal derived fromthe control signal therefore controlling the voltage levels at the firstand second supply output terminals of the variable voltage power supply.

The variable voltage power supply further may comprise a DC-DCconverter, such that the input voltage of the charge pump circuit isderived from the output of the DC-DC converter, and wherein the outputof the DC-DC converter depends on a signal derived from the controlsignal.

The variable voltage power supply may comprise a switch allowing theDC-DC converter to be bypassed and the input voltage of the charge pumpcircuit to be obtained directly from the input of the DC-DC converter.

The variable voltage power supply further may comprise a linearregulator between the output of the DC-DC converter and the input of thecharge pump circuit.

The variable voltage power supply may be arranged to vary the supplyvoltage of the output stage between a plurality of discrete voltagelevels in response to the control signal. Alternatively the variablevoltage power supply may be arranged to vary the supply voltage of theoutput stage in a substantially continuous and corresponding manner inresponse to the control signal.

The output voltage of the variable voltage power supply minus apredetermined offset may be substantially proportional to the gain inthe circuit. The predetermined offset may be substantially constant andindependent of the control signal. Alternatively the predeterminedoffset may be dependent on the control signal.

The variable voltage power supply may be arranged to vary the supplyvoltage such that reductions in the amplitude of the output signalcaused by a variation of the control signal may be not matched byincreases in voltage drop within the output stage, or the variablevoltage power supply may be arranged to vary the supply voltage suchthat reductions in the amplitude of the output signal caused by avariation of the control signal may be not matched by increases in powerloss within the output stage.

The amplifier circuit may comprise a linear amplifier, for example aclass A or class AB amplifier.

The gain controller may comprise a variable gain amplifier that may bein the signal path prior to the output stage and that may be responsiveto the control signal.

The gain controller may be comprised in the output stage, the controlsignal being arranged to control the signal amplitude at the main outputterminal by acting directly on the output stage, or the gain controllermay include the output stage, the control signal being arranged tocontrol attenuation of a signal fed back from the main output terminalto an input of the output stage.

The amplifying circuit may be of a type adapted for the amplification ofaudio signals, wherein the control signal may be a volume controlsignal.

The invention also provides for an audio apparatus, portable audioapparatus, communications apparatus, in-car audio apparatus or headphoneamplifier incorporating an amplifier circuit or an output amplifierapparatus as described above.

The invention also provides for electronic apparatus comprising anoutput transducer and an amplifier circuit or an output amplifierapparatus as described above having its output terminal connected todrive the output transducer as the load.

The invention further provides for an RF transmitter apparatuscomprising an amplifier circuit or an output amplifier apparatus asdescribed above having its output stage adapted to drive an antenna asthe load.

The invention further provides for a line driver for driving a signalthrough a transmission line, the line driver incorporating the signalamplifying apparatus as described above adapted for driving atransmission line as the load. The line driver may comprise part of amodem device further comprising a modulator, demodulator and controller.

The invention also provides for a method of amplifying an input signalto generate a gain controlled output signal, the method comprising:

generating a split-rail supply voltage from a single input supplyreceived across a supply input terminal and a common terminal, thesplit-rail supply being output at first and second supply outputterminals connected to the common terminal via respective first andsecond reservoir capacitors, the method comprising connecting a flyingcapacitor between different ones of the supply terminals in a sequenceof states, so as to transfer packets of charge repeatedly from the inputsupply to the reservoir capacitors via the flying capacitor and therebyto generate the split rail supply with positive and negative outputvoltages together spanning a voltage approximately equal to the voltageof the input supply, and centered on the voltage at the common terminal;the method further comprising:

applying the split-rail supply voltage to an output stage of anamplifier circuit;

receiving an input signal on a first amplifier input terminal of theamplifier circuit;

receiving a control signal on a second amplifier input terminal of theamplifier circuit;

applying a gain to the input signal in response to the control signal toproduce the gain controlled output signal at an amplifier outputterminal of the output stage of the amplifier circuit wherein the gainis independent of the supply voltage of the output stage; and

varying the split-rail voltage supply applied to the output stage inresponse to the control signal.

BRIEF DESCRIPTION OF THE DRAWINGS

Example embodiments of the invention are described hereinafter withreference to the accompanying drawings in which:

FIG. 1 shows a prior art amplification circuit;

FIG. 2 a shows apparatus according to a first embodiment of theinvention;

FIG. 2 b illustrates a signal flow diagram of the first embodimentillustrated in FIG. 2 a;

FIG. 3 shows waveforms associated with the embodiment of FIG. 2 a;

FIGS. 4(a) to 4(c) show waveform relationships associated with theembodiment of FIG. 2 a;

FIG. 5 a shows a Level Shifting Charge Pump circuit suitable for use inthe variable voltage power supply in any embodiment of the invention;

FIG. 5 b shows the same circuit as FIG. 4 a with detail of the switcharray shown;

FIGS. 6 a and 6 b show, respectively, the circuit with the switch arrayin a first state and an equivalent circuit of this state;

FIGS. 7 a and 7 b show, respectively, the circuit with the switch arrayin a second state and an equivalent circuit of this state;

FIGS. 8 a and 8 b show, respectively, the circuit with the switch arrayin a third state and an equivalent circuit of this state;

FIG. 9 is a timing diagram showing three switch control signals for thecircuit of FIGS. 1 and 2 operating in a first main mode (Mode 1);

FIG. 10 shows a Dual Mode Charge Pump circuit suitable for use in thevariable voltage power supply in any embodiment of the invention;

FIGS. 11 a and 11 b show, respectively, the circuit with the switcharray in a sixth state and an equivalent circuit of this state;

FIGS. 12 a and 12 b show, respectively, the circuit with the switcharray again in the second state and an equivalent circuit of this state;

FIG. 13 is a timing diagram showing control signals in a first variantof a second main mode of operation (Mode 2(a));

FIGS. 14 a and 14 b show, respectively, the circuit with the switcharray in a seventh state and an equivalent circuit of this state;

FIGS. 15, 16 and 17 are timing diagrams showing switch control signalsin second, third and fourth variants of the second main mode ofoperation (Mode 2(b), 2(c), 2(d) respectively);

FIG. 18 shows a variation on the circuit of FIG. 5 , operable in aclosed loop configuration;

FIG. 19 shows a variable voltage power supply of a type suitable for anyof the novel amplifiers disclosed herein whereby a number of differentinput voltage values may be selected as an input voltage to any of theLevel Shifting/Dual Mode Charge Pumps disclosed herein;

FIG. 20 shows a variable voltage power supply of a type suitable for anyof the novel amplifiers disclosed herein;

FIGS. 21 a-21 e show apparatus according to an embodiment of theinvention with alternatives;

FIGS. 22 a-22 c shows apparatus according at an embodiment of theinvention with alternatives;

FIGS. 23 a and 23 b show two alternative apparatus according to anembodiment of the invention;

FIG. 24 shows schematically a first system using an embodiment of theinvention and

FIG. 25 shows schematically a second system using an embodiment of theinvention.

DETAILED DESCRIPTION

Example embodiments of circuitry, apparatus and methods described belowprimarily concern audio applications. However, it will be appreciated bythose skilled in the art that other applications to which the presentinvention is equally applicable are possible and a few such applicableapplications are herein described and illustrated.

Basic Amplifier Design

FIG. 2 a illustrates an embodiment of a novel amplifier 100 that hasbeen designed to improve the efficiency over devices such as theamplifier 10 described above.

In this particular embodiment, the amplifier 100 comprises: the gaincontroller 20; the output stage 40 and the power supply 60 as describedabove. However, amplifier 100 differentiates itself in a first respectfrom amplifier 10 in that it includes a variable voltage second supply80 in addition to the fixed voltage first power supply 60.

The gain controller 20 receives an audio input signal S1 and a gain,e.g. a volume, control signal S2′. Amplifier 100 differentiates itselfin a second respect from amplifier 10 in that, the control signal S2′has a dual purpose. One such purpose is to control the gain controller20. The controller 20 outputs a gain controlled signal S3, used as aninput signal into the output stage 40 which in turn outputs an outputsignal S4′ that is used to drive a load 70, such as a speaker forexample.

The output stage 40 is supplied by the variable voltage power supply 80which is controlled in response to the control signal S2′. Therefore,the single control signal, S2′, has the dual purpose of: (1) controllingthe gain controller 20; and (2) controlling the variable voltage powersupply 80. It should be noted that the output stage 40 is independent orsubstantially independent (ignoring power supply rejection issues andthe like) of the variable voltage power supply 80. The operationalefficiency of the amplifier 100 is affected by the voltage of itssupplies and in particular the voltage supplied to the output, i.e.power, stage 40. Therefore, the single control signal, S2′, controlsboth the gain of the amplifier 100 and its operational efficiency, aswill be described in more detail below.

Moreover, the variable voltage power supply 80 is operatively arrangedin such a way, that the output stage 40 supplies voltages +Vout and−Vout which are varied sufficiently enough in response to the controlsignal S2′ to avoid clipping of the output signal S4′. This will bedescribed in more detail below,

The variable voltage power supply 80 receives a supply voltage from apower source (not illustrated), such as, but not necessarily, a fixedvoltage first power supply 60. The variable voltage power supply 80 isof a type which includes either a “Level Shifting Charge Pump” or a“Dual Mode Charge Pump” circuit as described later. This charge pumpmay, in turn, receive its input voltage from a variable voltage DC-DCconverter (such as a Buck Converter) either directly or via a linearregulator such as a Low drop out regulator. This allows the charge pumpoutputs to be varied as required by controlling the DC-DC converter andtherefore the input to the charge pump.

By way of one possible illustrative example of how the amplifier 100 maybe used and controlled let us assume that the amplifier 100 is an audioamplifier for amplifying an audio input signal S1 wherein: the variablevoltage power supply 80 is a Level Shifting Charge Pump circuit; thegain controller 20 and output stage 40 are linear amplifiers, such asclass AB amplifiers; the control signal S2′ is a volume control signal;and the load 70 is a speaker.

The control signal S2′ controls the overall gain of the amplifier 100 inorder to change the output volume of the speaker 70. The output volumemay be changed in a conventional manner wherein the output volume, i.e.the amplitude of the output signal S4′, is varied in response to avolume controller (not illustrated), such as a potentiometer, beingmanipulated by a user. Therefore, the input signal S1 is scaled by afactor determined by the gain of the amplifier 100 which is controlledin response to the volume control signal S2′.

However, according to the novel amplifier 100, the volume control signalS2′ also controls the variable voltage power supply 80. Therefore, thevariable voltage power supply 80 produces both positive and negativeground reference supply voltages, respectively +Vout and −Vout, thatvary in response to the volume control signal S2′.

It should be noted that In order to prevent the output signal S4′ fromever clipping, i.e. distorting, the amplifier 100 should be designed andcontrolled such that:Vout=|VS4′|+Vx  (Equation 1)

-   -   where;        -   Vout is the magnitude of output voltage of the variable            voltage power supply 80;        -   |VS4′| is the maximum voltage amplitude of the output signal            S4′; and        -   Vx is the headroom voltage between output signal S4′ and            supply voltage Vout that is required by the amplifier output            stage 40 to avoid the output signal S4′ clipping;

andVS4′=VS1max×G  (Equation 2)

-   -   where:        -   VS1max is a predetermined maximum permissible voltage            amplitude of the input signal S; and        -   G is the gain of the amplifier 100, as determined by the            respective gains of the controller 20 and the output stage            40.

VS1max will generally be predetermined from the design specification ofthe system, in terms of the maximum input voltage permissible toguarantee avoidance of clipping or to guarantee some other signaldistortion specification. In some cases, an application may receivesignals larger than the anticipated, i.e. designed, maximum signal andas a result, the output signal may clip or give extra distortion, butperformance under such overload conditions is not important. In somecases, for example where the input signal is derived from a digitalsource or is output from a DAC, there may be a well-defined maximumsignal level, set by the word-length or the full-scale reference voltageof the DAC, which the input signal can never exceed.

From Equations 1 & 2, it can be seen that the output voltage Vout of thevariable voltage power supply 80 is preferably linearly dependent on thegain G of the amplifier 100 for a given maximum input signal VS1max.

It can be appreciated from the above description that when the volume,i.e. gain, control signal S2′ is increased, the output signal S4′amplitude, and hence volume, increases as a result of the increased gainG of the amplifier 100. At the same time, the volume control signal S2′acts upon the variable voltage power supply 80 and changes its outputvoltages +Vout and −Vout accordingly in response to the control signalS2′. The way that the variable voltage power supply 80 changes theoutput voltages will become apparent later.

It should be noted that the headroom voltage Vx is preferably kept to aminimum, for a particular design embodiment, so as to minimise the powerloss in the amplifier and help maintain overall efficiency.

The variable voltage power supply 80 may be designed such that itsoutput voltages +Vout and −Vout change substantially continuously withthe control signal S2′. This may include the possibility of a digitalcontrol (not illustrated) with fine resolution. Alternatively, thevariable voltage power supply 80 may be designed such that its outputvoltages +Vout and −Vout change between a plurality of discrete voltagelevels as the control signal S2′ changes.

FIG. 2 b illustrates a signal flow diagram of the embodiment illustratedin FIG. 2 a.

From FIG. 2 b it can be seen that: signal Sc is a function of signals Saand Sb; signal Sd is a function of signal Sb; and signal Se is afunction of only signal Sc since signal Se is independent, orsubstantially independent, of signal Sd, wherein: signal Sa representsthe input signal S1; signal Sb represents the control signal S2′; signalSc represents the gain controlled signal S3; signal Sd represents thevoltage signal Vout; and signal Se represents the output signal S4′.

FIG. 3 illustrates a waveform plot of voltage against time for thearrangement of FIG. 2 a over a time span during which the volume controlsignal S2′ is reduced.

Period T1 of FIG. 3 represents the amplifier 100 when its control signalS2′ is set to its maximum value. It should be noted that during thisperiod the efficiency of both the respective amplifiers 10 and 100 ofFIGS. 1 and 2 would be the same, or substantially the same for the samesignal conditions, since their respective supply voltages +/−V1 and+/−Vout are equal.

Referring to Period T1 in conjunction with FIG. 2 a and considering justthe positive excursion of the output signal S4′ (since the same equallyapplies to the negative excursion), the power delivered to the load 70at the peak voltage VS4′max1 of the output signal S4′ is the product ofthe load current IL1 (not illustrated) and VS4′max1: whereIL1=VS4′max1/RL and RL is the resistance of the load 70. The voltageVS4′max1 is specified to allow a certain headroom voltage Vx between thepeak output signal voltage VS4′max1 and the supply voltage +Voutmax inorder for the correct operation of the output stage 40, whereVx=+Voutmax−VS4max1. The power dissipated by having this headroomvoltage Vx is wasted power PW1 which is given by the product of the loadcurrent IL1 and Vx. This power PW1 serves no purpose other than toensure the correct operation, i.e. it avoids distortion through signalclipping, of the output stage 40.

It can be seen that during period T1, where S4′ is at or near themaximum signal level that either amplifier 10, 100 can comfortably copewith, the amplifier 100 of FIG. 2 a operates in substantially the sameway and is therefore no more efficient than the amplifier 10 illustratedin FIG. 1 since under the conditions of period T1 the respectiveamplifiers 10 and 100, supply voltages +/−V1 and +/−Vout are the same.

Period T2 of FIG. 3 represents the amplifier 100 when its control signalS2′ is set to a value between its maximum and minimum values andtherefore output signal S4′ has a smaller amplitude than during periodT1. Unlike for the period T1, the respective efficiencies of therespective amplifiers 10 and 100 of FIGS. 1 and 2 are substantiallydifferent since the output voltage of amplifier 10 of FIG. 1 remains, asalways, at its fixed level +/−V1, as indicated by the dash-dot lines,whereas the dynamic output voltage +/−Vout of the variable voltage powersupply 80 has been adjusted, by the control signal S2′, to a new level+/−Voutbet. It can been seen that during this period T2 the efficiencyof the amplifier 100 of FIG. 2 a has been improved quite substantiallyover that associated with the amplifier 10 of FIG. 1 as can be deducedfrom comparing the amplitudes of the voltages Vx and Va during thisperiod T2.

This improvement in efficiency can be seen by referring to Period T2 inconjunction with FIG. 2 a , over the positive excursion. The powerdelivered to the load 70 at the peak voltage VS4′max2 is the product ofthe load current IL2 (not illustrated) and VS4′max2: whereIL2=VS4′max2/RL. Again the power dissipated by having this headroomvoltage Vx is wasted power. However, the amplifier 100 during period T2operates differently to the amplifier 10 of FIG. 1 , in having supplyrails at +/−Voutbet, while the supply voltages of amplifier 10 are fixedat +/−V1. Therefore it can be seen that the power PW2 saved by theamplifier 100, over amplifier 10, is given by the product of the loadcurrent IL2 and the voltage Va, where Va=V1−Voutbet.

Period T3 of FIG. 3 represents the amplifier 100 when its volume controlsignal S2′ is set to its minimum value, such that the output signallevel is very low but possibly still audible. Again, unlike for periodT1, during this period T2 the respective efficiencies of the respectiveamplifiers 10 and 100 of FIGS. 1 and 2 are substantially different sincethe output voltage of amplifier 10 of FIG. 1 remains at its fixed levels+/−V1, as indicated by the dash-dot lines, whereas the dynamic outputvoltage +/−Vout of the variable voltage power supply 80 has beenadjusted, by the control signal S2′, to a new level +/−Voutmin. It canbeen seen that during this period T3 the efficiency of the amplifier 100of FIG. 2 a has been improved substantially over that associated withthe amplifier 10 of FIG. 1 as can be deduced from comparing theamplitudes of the voltages Vx and Vb during this period T3.

The improvement in efficiency is again illustrated by referring toPeriod T3 in conjunction with FIG. 2 a over the positive excursion. Thepower delivered to the load 70 at the peak voltage VS4′max3 is theproduct of the load current IL3 (not illustrated) and VS4′max3, whereIL3=VS4′max3/RL. As before, the power dissipated by having this headroomvoltage Vx is wasted power. However, the amplifier 100 during period T3operates differently to the amplifier 10 of FIG. 1 , in having supplyrails at +/−Voutmin, while the supply voltages of amplifier 10 are fixedat +/−V1. Therefore it can be seen that the power PW3 saved by theamplifier 100, over amplifier 10, is given by the product of the loadcurrent IL3 and the voltage Vb, where Vb=V1−Voutmin.

Therefore, in general, for periods T2 and T3 the instantaneous power PWisaved by amplifier 100, over amplifier 10, is the product of theinstantaneous load current ILi and the voltage difference between V1 andVou.; Over a period of time the average saved power PWa is the productof the average load current ILa and the voltage difference between V1and Vout.

Therefore, as can be deduced from FIG. 3 in conjunction with FIG. 2 a ,by adapting i.e. dynamically changing, the supply voltage +/−Vout of theoutput stage 40 in response to the gain control signal (and thereforeeffectively the maximum swing of the output signal S4′, preferablyallowing for a headroom voltage Vx), the efficiency of the output stage40 and amplifier 100 is improved over that associated with the amplifier10 of FIG. 1 .

It should be noted that in FIG. 3 , Vx is illustrated as being constantor substantially constant, however it to may also be possible to furtherimprove the efficiency by allowing Vx to vary with the control signal.For instance a particular output stage may require less headroom whenoutputting lower output currents, so Vx and hence Vout can be reduced atcontrol input settings related to lower gain settings.

It can therefore be seen that the amplifier 100 of FIG. 2 aadvantageously reduces losses and improves efficiency when the controlsignal S2′ is used to control the variable voltage power supply 80 so asto control the variation of the supply voltage +Vout/−Vout supplied tothe ‘power amplifying’ output stage 40 in addition to controlling thegain G of the amplifier 100.

FIGS. 4(a)-4(c) illustrate example relationships between the controlsignal S2′ and the supply voltage to the output stage +/−Vout as thecontrol signal S2′ is varied and used for controlling the variablevoltage power supply 80 in two different modes.

FIG. 4(a) is an example illustration of the control signal S2′ as it islinearly varied from its minimum value to its maximum value, and thensometime later, back down to its minimum value.

FIG. 4(b) illustrates the output voltage +/−Vout variation of thevariable voltage power supply 80 for the case where this power supply isdesigned to only output a plurality of discrete output voltages. Thismay give a simpler and hence cheaper structure for the variable voltagepower supply 80. It may be undesirable from an overall system efficiencypoint of view to generate intermediate voltages. In this situation, tocontrol the output voltage as a function of the control signal S2′, aset of threshold levels is defined as indicated by the referencesTr1-Tr3 in FIG. 4(a). Since Vout must always guarantee the headroom Vxabove the anticipated maximum output signal swing, yet there are only afew possible levels for Vout, Vout will generally be somewhat largerthan the minimum value possible. The dashed line in FIG. 4(b) thattracks the voltages represents the same waveforms as presented in FIG.4(c) and illustrates the “waste” of voltage, i.e. the inefficiency. Sowhile this ‘discrete voltage level’ mode is more efficient than thatassociated with the amplifier 10, it is not as efficient as the modeassociated with FIG. 4(c).

FIG. 4(c) illustrates an example variation of the output voltage +/−Voutof the variable voltage power supply 80, as a function of the controlsignal S2′, when the control signal S2′ controls the variable voltagepower supply 80 such that a variation in the control signal S2′ causes asubstantially continuous and corresponding variation in the outputvoltage +/−Vout. In this case the variation in the output voltage+/−Vout follows a variation in the control signal S2′. Vout iscontrolled so that it tracks the maximum anticipated output swing, withextra headroom Vx.

Many modern amplifiers may have the gain digitally controlled, in whichcase there will still be discrete levels of the control signal, but somany of them (say 256 for an 8-bit control word), that the resultantsupply voltage waveform +/−Vout will substantially be similar to that ofFIG. 4 c . In such an embodiment, the loss of efficiency, due to thefinite resolution in supply voltage +/−Vout, will be small and thesupply voltage +/−Vout may be regarded as being varied in asubstantially continuous manner.

Also illustrated, for comparison, in FIGS. 4 a-4 c is a dash-dot linethat represents the fixed voltage level +/−V1 associated with thearrangement of FIG. 1 .

Variable Voltage Power Supply Design

The variable voltage power supply 80 will now be described. As mentionedpreviously this power supply includes a charge pump of a novel typereferred to as a “Level Shifting Charge Pump” (LSCP) or a variation onthis referred to as a “Dual Mode Charge Pump”. These charge pumpcircuits address the problems of conventional charge pump circuits, suchas Inverting Charge Pumps, namely, that they can only generate outputvoltages that have a rail-to-rail magnitude greater than the inputvoltage. This can be disadvantageous in certain applications, as it maynot allow the circuitry being supplied to run efficiently, for examplewhen such an Inverting Charge Pump circuit is being used to powercircuitry that amplifies a signal with a maximum amplitude much smallerthan the amplifier circuitry's power supply +/−VDD. This means that suchan inverting charge pump, should it be used in the Variable VoltagePower Supply 80 for the novel amplifier 100, may be inefficient atparticularly low volumes where the appropriate output level of thecharge pump for the volume set is somewhat less than its lowest possibleinput level. Furthermore, should the charge pump receive its input froma DC-DC converter, there would be significant losses in this DC-DCconverter should it have to input lower voltages to the charge pump muchof the time.

FIG. 5 a is a block diagram of a novel inverting charge pump circuit,which we shall call a “Level Shifting Charge Pump” (LSCP) 400. In thiscircuit there are two reservoir capacitors CR1 and CR2, a flyingcapacitor Cf and a switch array 410 controlled by a switch controller420 (which may be software or hardware implemented) arranged as shown.In comparison to a conventional Inverting Charge Pump, it is notablethat reservoir capacitor CR1 is not connected directly to the inputsupply voltage VDD, but only via the switch array 410.

It should be noted that LSCP 400 is configured as an open-loopcharge-pump. Therefore, LSCP 400 relies on the respective loads (notillustrated) connected across each output N12-N11, N13-N11 remainingwithin predetermined constraints. The LSCP 400 outputs two voltagesVout+, Vout− that are referenced to a common voltage supply (node N11).Connected to the outputs Vout+, Vout−, N11, and shown for illustrationonly, is a load 450. In reality this load 450 may be wholly or partlylocated on the same chip as the power supply, or alternatively it may belocated off-chip.

LSCP 400 operates such that, for an input voltage +VDD, the LSCP 400generates outputs each of a magnitude which is a half of the inputvoltage VDD. In other words, the output voltages generated in this firstmode are nominally of magnitude +VDD/2 and −VDD/2. When lightly loaded,these levels will, in reality, be +/−(VDD/2−Iload.Rload), where Iloadequals the load current and Rload equals the load resistance. It shouldbe noted that, in this case, the magnitude (VDD) of output voltageacross nodes N12 & N13 is the same, or is substantially the same, asthat of the input voltage (VDD) across nodes N10 & N11, but shifted.

This particular form of charge pump has significant advantages overknown circuits, in particular because of the ability to generated areduced, bipolar supply using only a single flying capacitor. Priorcircuits for generating reduced output voltages requires additionalflying capacitors. The flying capacitor and reservoir capacitors areoften of a size that they need to be located off-chip, and soeliminating one capacitor and two IC pins is highly beneficial.

FIG. 5 b shows more internal detail of the LSCP 400 and, in particular,detail of the switch array 410 is shown. The switch array 410 comprisessix switches S1-S6 each controlled by corresponding control signalCS1-CS6 from the switch controller 420. The switches are arranged suchthat first switch S1 is connected between the positive plate of theflying capacitor Cf and the input voltage node N10, the second switch S2is between the positive plate of the flying capacitor and first outputnode N12, the third switch S3 is between the positive plate of theflying capacitor and common terminal N11, the fourth switch S4 isbetween the negative plate of the flying capacitor and first output nodeN12, the fifth switch S5 is between the negative plate of the flyingcapacitor and common terminal N11 and the sixth switch S6 is between thenegative plate of the flying capacitor and second output terminal N13.Optionally, there may be provided a seventh switch S7 (shown dotted onFIG. 10 ), connected between the input voltage source (node N10) andfirst output node N12. These switches are the ones appropriate tooperate as described herein. The provision of further switches to enableother modes of operation is of course not excluded.

It should be noted that the switches can be implemented in a number ofdifferent ways (for example, MOS transistor switches or MOS transmissiongate switches) depending upon, for example, an integrated circuit'sprocess technology or the input and output voltage requirements. Theselection of appropriate implementations is well within the capabilityof the skilled reader.

The LSCP 400 has three basic states of operation repeated inhigh-frequency cycles of three phases, which may be referred to as P1,P2, P3.

FIGS. 6 a and 6 b show the switch array 410 operating in a first state,“State 1”. Referring to FIG. 6 a , switches S1 and S4 are closed suchthat capacitors Cf and CR1 are connected in series with each other andin parallel with the input voltage +VDD. Therefore, capacitors Cf andCR1 share the input voltage +VDD that is applied across them. FIG. 6 bshows an equivalent circuit for the State 1 operation with voltage +VDDeffectively applied across nodes N10 & N11.

It is preferable in order to obtain symmetrical, opposite polarity,output voltages, that the values of capacitors Cf and CR1 are equal suchthat each capacitor Cf, CR1 changes voltage by an equal increment whenconnected in series across a voltage source. If both capacitors areinitially discharged, or indeed previously charged to any equalvoltages, they will end up each with a voltage equal to half the appliedvoltage source, in this case one half of the input voltage VDD.

FIGS. 7 a and 7 b show the switch array 410 operating in a second state,“state 2”. Referring to FIG. 7 a , switches S3 and S6 are closed suchthat capacitors Cf and CR2 are connected in parallel with each other andbetween nodes N11 and N13. Therefore, the voltage across capacitor Cfequalises with that across capacitor CR2. Over a plurality of cycles,the voltages across the capacitors Cf and CR2 will converge to a voltageVDD/2. FIG. 7 b shows an equivalent circuit for this state 2 condition.

It should be noted that the value of reservoir capacitor CR2 does notnecessarily need to be the same as that of flying capacitor Cf. Ifcapacitor CR2 is much larger than capacitor Cf, it will require morestate sequences to charge up to or close to VDD/2. The value ofreservoir capacitor CR2 should be chosen depending upon expected loadconditions and required operating frequency and output ripple tolerance.

Over a plurality of cycles alternating only States 1 and 2, the voltagesacross the capacitors Cf and CR2 would, under ideal conditions, convergeto a voltage +/−VDD/2. However, the presence of a significant load onthe LSCP's 400 output terminals will result in a respective voltagedroop in Vout+, Vout− away from +/−VDD. If the load is symmetric, andthere is equal current magnitude on both Vout+ and Vout−, then thesymmetry of the system will result in both outputs drooping by the sameamount.

However, if for example there is a significant load on Vout+ but no loador a light load on Vout−, then the voltage across capacitor CR1 willreduce. This will result in a larger voltage across capacitor Cf at theend of State 1 which will then be applied to capacitor CR2 in State 2.If only States 1 and 2 were used, the flying capacitor Cf would then beconnected in series with capacitor CR1 in State 1 but still having alarger voltage across it, even initially. Therefore, voltages Vout+ andVout− will both tend to droop negatively, that is to say that the commonmode is not controlled.

To avoid this effect, a third state, State 3, is introduced and States 1to 3 are repeated in Phases 1 to 3 over successive cycles. FIGS. 8 a and8 b show the switch array 410 operating in this state 3 operation.Referring to FIG. 8 a , in state 3, switches S2 and S5 are closed suchthat capacitors Cf and CR1 are connected in parallel with each other andbetween nodes N11 and N12. Therefore, both capacitors Cf and CR1 becomecharged up to an equal voltage, despite any difference between of theirprevious voltages. In steady state (after many cycles) this becomesapproximately VDD/2. FIG. 8 b shows an equivalent circuit for this State3 condition.

The circuit, therefore ends State 3 with equalised voltages, after whichit returns to State 1. Consequently the circuit will, in principle,enter Phase 1 of the next cycle in State 1 with Vout+=+VDD/2, dependingupon load conditions and switching sequence.

In States 2 and 3, the voltages across the various capacitors that areconnected in parallel may not actually, in practice, completely equalisein a single sequence, particularly if the switching frequency is high,relative to the LSCP's R-C time constant. Rather, in each sequence ofstates a contribution of charge will be passed from capacitor tocapacitor. This contribution will bring each output voltage to thedesired level under zero, or low, load conditions. Under higher loadconditions, the output reservoir capacitors CR1, CR2 will typicallyachieve a lower voltage (with some ripple). The size of each of thecapacitors needs simply to be designed such that the reduction of commonmode drift is within acceptable bands for all expected load conditions.Alternatively, or in addition, larger switches, with less on-resistance,could be employed.

FIG. 9 illustrates the non-overlapping control signals (CS1-CS6) forcontrolling the switches (S1-S6) during the three states (1, 2 and 3) ofthe main operational embodiment. As discussed above, this representsonly one example out of many possibilities for the controlling sequence.

It should be appreciated that the open-loop sequencing of the abovethree states does not necessarily need to be observed. For example thestate sequences could be: 1, 2, 3, 1, 2, 3 . . . (as described above);or 1, 3, 2, 1, 3, 2 . . . ; or 1, 2, 1, 3, 1, 2, 1, 3. It should also beapparent that it is not necessary that the third state be used as oftenas the other two states, for instance a sequence of 1, 2, 1, 2, 1, 2, 3,1 . . . can be envisaged. It may even be envisaged to dispense with thethird state altogether, albeit only in the case of well-balanced loads,or with alternative schemes for common-mode stabilisation.

Other switching and sequencing scenarios exist. For example, in onealternative operational Mode 1 embodiment: State 1 could be replaced bya fourth state, “State 4” whereby switches S1 and S5 are closed (allother switches are open). In this state capacitor Cf charges up to inputvoltage +VDD. A fifth state, “State 5” would then operate with switchesS2 and S6 closed (all other switches open) such that flying capacitor Cfis connected across reservoir capacitors CR1 and CR2 in series (which,in this scenario, may be equal in capacitance). This particular exampleof an alternative switching and sequencing scenario has the drawbackthat there is no common-mode control and therefore such a switching andsequencing scenario would suffer from common-mode drift. However, thiscommon-mode drift can be “reset” by altering the switching sequence atappropriate intervals during the “normal” switching and sequencingcycle. These alterations can be predetermined, or initiated in responseto observed conditions.

It should be noted that the sizes of capacitors Cf, CR1, CR2, can beselected to meet the required ripple tolerances (versus size/cost) andconsequently the clock phase duration for each state need notnecessarily be of ratio 1:1:1.

While the above describes an embodiment wherein the LSCP generatesoutputs of +/−VDD/2, it will be understood by the skilled person thatthe above teaching could be used to obtain outputs of any fraction ofVDD by increasing the number of flying capacitors Cf and altering theswitch network accordingly. The relationship between output and input inthis case is Vout+/−=+/−VDD/(n+1) where n equals the number of flyingcapacitors Cf. It will also be appreciated that circuits with more thanone flying capacitor as described will still be capable of generatingoutputs of +/−VDD/2 as well as outputs for every intermediate integerdenominator between +/−VDD/2 and +/−VDD/(n+1) depending on its control.For example, a circuit with two flying capacitors can generate outputsof VDD/3 and VDD/2, one with three flying capacitors can generateoutputs of VDD/4, VDD/3 and VDD/2 and so on.

Obviously, in order to operate as a variable voltage power supply, theLSCP needs to have variable outputs. This may be achieved as describedin the above paragraph. It may also be achieved by having the inputvoltage +VDD alterable in any suitable way, one example beingillustrated below with reference to FIGS. 16 and 17 . Another way is bymaking the reference voltage alterable on a LSCP circuit operating in aclosed loop configuration as illustrated in FIG. 15 . Alternatively, orin addition to these methods, the circuit of FIG. 5 a can also be madecapable of dual mode operation depending on its controllingcircuitry/programming, each mode resulting in different output voltagelevels.

When the LSCP is configured to be operable in two modes the circuit willbe referred to as the Dual-Mode Charge Pump (DMCP). In this embodiment,there is provided a mode select circuit 430 within the control module420. This a mode select circuit 430, depending on an input controlsignal Ic, selects one of two switch controller circuits/programs 420 a,

FIG. 10 shows an alternative embodiment referred to as the Dual-ModeCharge Pump (DMCP) which is operable in two main modes. The charge-pump,in this example configured as an open-loop charge-pump, differs in thatthere is provided a control module 420 which, at least notionally,comprises mode select circuit 430 for deciding which of two controlfunctions 420 a, 420 b to use, thus determining which mode the DMCPoperates in. The mode select circuit 430 and the controllers 420 a, etc.are notional blocks in that they represent different behaviours of thecontrol module in implementing different operating modes of DMCP 400.They can be implemented by separate circuits as just described. Inpractice, they are just as likely to be implemented by a single circuitblock or sequencer with hardwired logic and/or sequencer codedetermining which behaviour is implemented at a given time. As alsodescribed below, where a given mode can be implemented in a range ofvariants, the designer may select variants which simplify the generationof the control signals, when all the different modes are consideredtogether.

Another optional difference between the LSCP and DMCP is that the switcharray 1100 now comprises seven switches S1 to S7. Switches S1 to S6 arearranged as before, while optional switch S7 is connected between theinput voltage source and first output node N12.

The DMCP's two main modes are a first mode (Mode 1) where it produces adual rail output of voltages +/−VDD/2, and a second mode (Mode 2) whereit produces a dual rail output of +/−VDD (+VDD again being the inputsource voltage level at node N10). As before, the circuit can alsoproduce outputs of any voltages up to these levels if arranged tooperate in a closed loop configuration.

Furthermore, in Mode 2, the circuit is operable in four sub-Modes,referred to as Modes 2a, 2b, 2c and 2d. Optional switch S7 is only usedin Modes 2c and 2d. Consequently, if switch S7 is not included, Mode 2is only operable in sub-Modes 2a and 2b.

In Mode 2a the DMCP has two basic states of operation. FIG. 11 a showsthe circuit operating in the first of these states, “State 6”. In thisstate, switches S1, S2 and S5 are closed (S3, S4 and S6 are open). Thisresults in capacitors Cf and CR1 being connected in parallel across theinput voltage +VDD, between nodes N10 & N11. Therefore, capacitors Cfand CR1 each store the input voltage +VDD. FIG. 11 b shows an equivalentcircuit for the State 6 operation.

FIG. 12 a shows the circuit operating in the second of these states,“State 2”, which is, in fact, the same state as state 2 in Mode 1,whereby switches S3 and S6 are closed (S1, S2, S4 and S5 are open).Therefore capacitors Cf and CR2 are connected in parallel between commonnode N11 and second output node N13. Therefore, capacitors Cf and CR2share their charge and Node 13 exhibits a voltage of −VDD after a numberof state sequences. FIG. 12 b shows an equivalent circuit for this State2 of operation.

FIG. 13 illustrates the non-overlapping control signals (CS1-CS3 &CS5-CS6) for controlling the switches (S1-S3 and S5-S6) during the twoalternating states of Mode 2(a). The sequence of states in this mode istherefore 6, 2, 6, 2, 6, . . . etc.

FIG. 14 a shows an additional state, “State 7”, which can be introducedinto this Mode 2(a) sequence to create a slightly differentimplementation, referred to now as Mode 2(b). In State 7, switches S1and S5 are closed (S2, S3, S4 and S6 are open). This state 7 connectsthe flying capacitor Cf across the input voltage +VDD. This state can befollowed by states 6 then 2 and then back to 7 etc. FIG. 14 b shows anequivalent circuit for this State 7 operation.

FIG. 15 illustrates the non-overlapping control signals (CS1-CS3 &CS5-CS7) for controlling the switches (S1-S3 and S5-S7) to generate arepeating sequence of the three states 7, 6, 2, 7, 6, 2, etc. . . . thatdefines Mode 2(b). Again, this represents only one example out of manypossibilities for the controlling sequence. The inclusion of State 7before State 6 is intended to isolate CR1 from the influence of CR2, andhence combat cross-regulation. On the other hand, the inclusion of State7 reduces the time available for charge transfer in the main States 2and 6, so that regulation as a whole may be improved if State 7 issimply omitted (Mode 2(a)). These are design choices.

Whichever pattern is chosen, one of the states may be used lessfrequently than the others (as was described above in relation to Mode1). For instance, if the loads on the two output nodes N12, N13 areunbalanced (either permanently or according to signal conditions), oneof the States 6 and 2 could be included less frequently than the other,as capacitor CR1 may need to be charged less frequently than capacitorCR2 or vice versa.

Modes 2(c) and (d) are further alternative modes of operation togenerate +/−VDD, which are possible when the DMCP is provided withswitch S7. This switch may used to replace the combined functionality ofswitches S1 and S2 for generating the positive output voltage at nodeN12 in applications where the high-side load, i.e. the load connectedbetween nodes N12 and N11, does not require a lot of current. This maybe where the load has a high input resistance as with a “Line Output”for a mixer for example. In such a case the size and the driverequirements of switch S7 can be reduced and modified compared to thoseof switches S1 and S2. Indeed, switch S7 can be constantly switched onduring operation in Mode 2(c) which has advantages in that there is lesspower required to drive the switches and switch S7 would not, in thecase of a MOS switch implementation, inject any charge into either nodesN10 or N12 due to its parasitic gate-drain and gate-source capacitances.It should also be noted that switch S1 is still required to operate soas to generate the negative output voltage −VDD. Still further, itshould be noted that switch S2 may be operated on an infrequent basis soas to also connect the flying capacitor Cf and high-side reservoircapacitor CR1 in parallel.

FIG. 16 illustrates the non-overlapping control signals (CS1-CS3 &CS5-CS7) for controlling the switches (S1-S3 and S5-S7) during the twoalternating states of Mode 2(c). Summarising Mode 2(c), therefore,switch S7 is permanently (or near permanently) closed. A modified State6 is used to charge the flying capacitor Cf and capacitor CR1 inparallel, this now being achieved by having switches S1, S5 and S7closed only. A modified State 2 is then used to transfer this charge tocapacitor CR2 via switch S3, S6 as before, but this time with capacitorCR1 still having voltage VDD across it due to S7 being closed.

FIG. 17 illustrates non-overlapping control signals (CS1-CS3 & CS5-CS7)for controlling the switches (S1-S3 and S5-S7) during three states in avariation of Mode 2(c) referred to as Mode 2(d). The difference relativeto Mode 2(c) is similar to the difference between Modes 2(a) and 2(b),in that an extra phase is inserted with the switches in State 7, whereinswitches S1 and S5 are closed (S2, S3, S4 and S6 are open; S7 can remainclosed throughout). Note that Mode 2(d) follows a sequence 7, 2, 6, 7,2, 6 . . . rather than 7, 6, 2. There is not necessarily any greatdifference in the effect of these modes, but the freedom to vary thesequence can simplify the control logic, as will be seen in thediscussion below.

TABLE 1 S1 S2 S3 S4 S5 S6 S7* State 1 1 0 0 1 0 0 0 State 2 0 0 1 0 0 1  1⁺⁺ State 3 0 1 0 0 1 0 0 State 4 1 0 0 0 1 0 0 State 5 0 1 0 0 0 1 0State 6 1 1 0 0 1 0 0 State 6⁺ 1 0 0 0 1 0 1 State 7 1 0 0 0 1 0 0 State7⁺⁺ 1 0 0 0 1 0 1 *if present ⁺Modes 2c and 2d ⁺⁺Mode 2d

Table 1 illustrates the switch (S1-S7) states for the seven statesdescribed above, with a “0” representing an open switch and a “1”representing a closed switch. Note that the switch network andcontroller do not need to implement all states 1 to 7, if only a subsetof the described modes will be used in a particular implementation.

Again, these four example sequences and seven or eight different statesof the switch network are not the only possibilities for the controllingsequence. Again, a number of different sequence implementations arepossible and some of these states may be used less frequently thanothers, depending on load.

FIG. 18 illustrates a similar LSCP/DMCP 900 circuit as illustrated inFIG. 4 or FIG. 10 except that the LSCP/DMCP 900 also includes twocomparators 910 a, 910 b for regulating the two output voltages.

It should be noted that LSCP/DMCP 900 represents a closed-loopLSCP/DMCP. Each of the comparators 910 a, 910 b compares theirrespective charge pump output voltages (Vout+, Vout−) with a respectivethreshold voltage (Vmin+, Vmin−) and outputs a respective charge signalCHCR1 and CHCR2. These charge signals CHCR1, CHCR2 are fed into theswitch control module 1420 to control the switch array 1410 causing theLSCP/DMCP to operate charging either the relevant reservoir capacitor.If either output voltage droops past its respective threshold, thecharge pump is enabled; otherwise the charge pump is temporarilystopped. This reduces the power consumed in switching the switches,especially in conditions of light load.

This scheme allows output voltages up to +/−VDD/2. It should be furthernoted that in this configuration, the LSCP/DMCP 900 may be used togenerate higher voltages, but with a drop in efficiency. In this case,the reference voltages (Vmin+/Vmin−) can be adjusted to adjust theoutput voltages accordingly. The flying capacitor Cf is charged up to+VDD (via switches S1 and S5) and then connected in parallel acrosseither reservoir capacitor CR1 (via switches S2, S5) or CR2 (viaswitches S3, S6) to raise their voltages to the levels set by thereference voltages. Such an operation increases the ripple voltages onthe reservoir capacitors CR1, CR2 but it also reduces switching losses.However, by scaling the reservoir capacitors CR1, CR2 relative to thecharging capacitor Cf, the ripple voltages can be reduced. It ispossible, therefore, for the gain control signal S2′ of FIG. 2 tocontrol the reference voltages (Vmin+/Vmin−) and therefore control theoutput voltages Vout+ and Vout− of the variable voltage power supply 80.

FIG. 19 illustrates a variable voltage power supply 80 utilising any ofthe novel Charge Pumps 400, 900 described above, wherein one of a numberof different input voltage values may be selected as an input voltage tothe LSCP/DMCP 400, 900. It shows an input selector 1000 having a numberof different voltage inputs (+Vin 1 to +Vin N), the actual input chosenbeing determined by control input Ic. The chosen voltage level thenserves as the input voltage VDD for the charge pump 400, 900.

FIG. 20 shows a more detailed variation of FIG. 19 and which may be usedas the variable voltage power supply 40 of the novel amplifier 100. Thisshows a buck converter fed by an input voltage +V1 from, for example, abattery. The buck converter also receives a control signal Cb. Theoutput of the buck converter is fed through a line regulator (in thiscase a low drop out regulator), before being input into any of theLSCP/DMCPs described above.

In use, the Buck Converter 1010 receives an input voltage +Va (5 v forexample) and outputs a lower voltage +Vb (3.2 for example). It ispreferable to pass the output voltage +Vb of the Buck Converter 1010through a Linear Regulator such as a Low Drop Out (LDO) 1020 regulatorbefore inputting the voltage from the Buck Converter 1010 into theLSCP/DMCP. The LDO 1020 receives the output voltage Vb from the BuckConverter 1010 and outputs a slightly lower voltage +Vin (3 v forexample) which constitutes the input voltage of the LSCP/DMCP 400, 900.

It is preferable to use the LDO 1020 since both the Buck Converter 1010and the LSCP/DMCP 400, 900 are switching regulators and it is preferableto clean up the switching effects relating to the output voltage +Voutof the Buck Converter 1010 before it is fed into the LSCP/DMCP 400, 900.

The output voltage +Vout of the Buck Converter 1010 can be adjusted viaan external control signal Cb, possibly by changing its duty cycle. Inthis way, the input to, and therefore the outputs from, the LSCP/DMCP400, 900 is/are controllable. When used as the Variable Voltage PowerSupply for any of the novel amplifiers disclosed herein, it is envisagedthat control signal Cb is, or is derived from, gain control signal S2′.Additionally, the output voltages of the LSCP/DMCP 400, 900 can beadjusted (independently) via an external control signal Cp.

An additional feature is a bypass switch 1030 that may be employed in asituation where there is a need to connect the input voltage +V1directly to the input of the LSCP/DMCP 400, 900. This feature is usefulwhere +V1 is supplied from a battery that has slowly discharged to avoltage level for which the Buck Converter 1010 cannot or cannotefficiently generate +Vout and hence +Vin.

Variations on the Basic Amplifier Design

FIG. 21 a illustrates a variant embodiment of the circuit of FIG. 2 a .This embodiment works in essentially the same manner as the embodimentdescribed in relation to FIG. 2 a above. The main difference in thisFIG. 21 a embodiment is that its output stage 45 combines the functionsof the gain controller 20 and output stage 40 of FIG. 2 a . Therefore,the FIG. 21 a output stage 45 receives the gain control signal S2′which, as will be described and illustrated below, may act on a feedbackloop within the output stage 45.

FIGS. 21 b-21 e illustrate a number of different methods in which theamplifier 100 of FIG. 21 a can be controlled by the control signal S2′.FIGS. 21 b-21 e illustrate non-exhaustive examples and many otherarrangements will be apparent to the skilled reader. Each of these FIGS.21 b-21 e shows detailed elements comprising, or included in, the outputstage 45 FIG. 21 a.

FIG. 21 b illustrates the output stage 45 comprising an amplifier 600and variable resistors R1 and R2 arranged as illustrated. The controlsignal S2′ acts to change the resistance of one or both resistors. Thegain G of the amplifier 600 is varied by varying the resistance ratio ofresistors R1 and R2. Consequently, only one of these resistors need bevariable and controlled by the control signal S2′. If both the resistorsare varied, then one resistor may be controlled by the control signalS2′ and the other may be controlled by a derivative signal S22, producedfor example by a signal inverter 610, so that when R1 increases, R2decreases, and vice versa. It should be noted that in this embodiment,resistors R1 and R2 represent a gain controller that is arranged tocontrol the gain G of the amplifier 600 applied along the signal path,the signal path extending from the input terminal of the amplifier 600to its output terminal, wherein the gain G is controlled in response tothe control signal S2′. Control signal S2′ may be a digital controlword, in which case S22 may be say the lower bits of the control word,while R1 may be controlled by the higher bits of the control word.

FIG. 21 c illustrates a variation of the output stage 45 illustrated inFIG. 21 b . FIG. 21 c illustrates an embodiment having respectiveresistor and switch arrangements, as illustrated, that represent theresistors R1 and R2. In this particular embodiment, the respectivecontrol signals DS2′, and its derivative DS22, are digital versions ofthe respective control signals S2′ and S22 illustrated in FIG. 21 b .Also, control signal may be a multibit control signal as indicated bythe line MB. FIG. 21 d illustrates a similar arrangement to thatillustrated in FIG. 21 c.

FIG. 21 e illustrates another variation of the output stage 45illustrated in FIG. 21 b.

FIG. 21 e illustrates a ganged potentiometer R2, R3. In this particularembodiment, the resistance of the ganged potentiometers R2 and R3 aredependent on the control signal S2′. The control signal S2′ controlsthese ganged potentiometers such that R2 is adjusted so as to vary thegain G of the amplifier 600 while R3 is adjusted so as to vary theoutput voltage Vout of the variable voltage supply 80. If S2′ controlsR2 to give a higher resistance, the output signal voltage swing willincrease. To allow for this, the wiper on R3 is moved to give a higherinput reference voltage into variable power supply 80.

FIG. 22 a illustrates a variation on FIG. 2 a wherein the gain controlis controlled digitally.

FIG. 22 a illustrates an amplifier 101 that comprises a digital signalprocessor (DSP) 500, such as a multiplier for example, and a digital toanalogue converter (DAC) 520, such as a resistor/switch network,inserted in place of the gain controller 20 illustrated in FIG. 2 a .The DSP 500 and DAC 520 form an input stage 530. The DSP 500 receives adigital input signal DS1 from a data source (not illustrated), (such asa solid-state memory or information carrier, such as a CD or DVD forexample), and a digital gain control signal DS2′. The gain controlsignal DS2′ acts upon the DSP 500 and as a result DSP 500 varies itsdigital input signal DS1 such that it outputs a gain controlled digitaloutput signal DS1′. The DAC 520 receives the gain controlled digitalsignal DS1′ and outputs a corresponding gain controlled analogue signalAS1 which is processed in the same manner as described above inconnection with FIG. 2 a.

In this FIG. 22 a , the DAC is driven by a single supply 60 while theoutput is driven from the dual, i.e. split, variable voltage powersupply 80′, with a level shifter 30 required to translate signal AS1 atquiescent voltage +V1/2 to a ground-referenced signal AS1′, but no levelshifter is required between the ground-referenced output S4′ and thegrounded load 70.

In a further variation, the digital control signal DS2′ may act directlyon the structure of the DAC 520 rather than actually modulate a voltage,by, for example, selecting the size of a capacitor periodicallyconnected to a fixed DAC full-scale reference, to scale a charge used torepresent DAC full-scale signal, or by selecting the size of aresistance connected to a fixed DAC full-scale reference to scale acurrent used to represent a DAC full-scale signal, rather than bydirectly modulating a (decoupled) reference voltage.

It should be noted that the variable voltage power supply 80′ in thisparticular embodiment should be designed to be controlled by a digitalcontrol signal DS2′ as opposed to an analogue control signal. The designof such a digitally controlled variable voltage power 80′ supply will bereadily appreciated and facilitated by those skilled in the art.

FIG. 22 b illustrates a variation of the digital control andmanipulation as performed by the DSP 500 and DAC 520 in FIG. 22 a.

FIG. 22 b illustrates the first DAC 520 as directly receiving thedigital input signal DS1 from a data source (not illustrated) andoutputting the gain controlled analogue signal AS1′. A second DAC 525replaces the DSP 500 and receives the digital control signal DS2′. TheDACs 520 and 525 form an alternative input stage 530′. This second DAC525 outputs an analogue gain control signal AS2′ that is used to controlthe first DAC 520. For example AS2′ may be used as the full-scalereference voltage for the DAC, so the output for a given digital inputword (DS1) will scale directly with the reference voltage i.e. the gaincontrol signal AS2′.

It should be noted that either the digital gain control signal DS2′ orits derived analogue equivalent AS2′ may be used to control anappropriately arranged variable voltage power supply.

FIG. 22 c illustrates an embodiment wherein the gain control ofamplifier 101 may act at multiple points in its signal path betweensignal input and output. An input digital signal DS1 is multiplied inDSP 500, the resultant scaled digital signal DS1′ is input to eitherinput stage 530 or its alternative 530 which scales the signal DS1′(perhaps as described above) to give an analogue signal AS1 which isthen scaled by a gain controller 20 to give a signal AS2 which is thenlevel shifted by a level shifter 30 to give a signal AS2′, which is thenfurther scaled within a variable gain output amplifier 45. Each of theelements 500, 530/30′, 20, 45 receives, from a controller block 700, arespective gain control signal, as illustrated, according to an overallinput gain control signal DS2′. Signal DS2′ is also used to derive theappropriate power supply control signal to feed into variable powersupply 80′. It would also be possible for the gain control signal DS2′to be a multibit control signal (as illustrated) comprising individualwords to control each gain block, and for controller 700 to calculatethe appropriate power supply control signal, according to a calculatedcascaded gain. The controller 700 may be implemented by means of alook-up table, such an implementation be readily understood by thoseskilled in the art.

FIGS. 21 and 22 illustrate that the actual gain control may act at anypoint or multiple points, in the amplifier's (100, 101) signal pathbetween signal input and output, whether it be in the digital oranalogue domain, and preceding or combined with the output stage 40.

FIG. 23 a illustrates a variation on the embodiment of FIG. 2 a for, inthe case of audio applications, stereo systems.

Dual input signals S11, S12 are fed into a gain unit 220 that comprise again controller 20 (not illustrated) and possibly a level shifter 30(not illustrated) for each of the input signals S11 and S12 driven froma fixed supply 60 (not illustrated). The gain unit 220 outputsrespective gain controlled signals S31′, S32′. The two gain controllers20 (not illustrated) are controlled by a common gain or volume or levelcontrol signal S2′. The respective gain controlled signals S31′, S32′are fed into respective output stages 401, 402, which output respectiveoutput signals S41′, S42′ which are then fed into left and rightspeakers (not illustrated) or a stereo headphone (not illustrated). Thepower unit 200 contains the power supplies 60 and 80 as illustrated inFIG. 2 a.

The headphones referred to above may be either physically connected, bymeans of electrical wires, to the amplifier 102 or they may not, inwhich case the headphones may, for example, receive the signals S11′ andS12′ via, for example, infrared or RF signals. In either of theseheadphone arrangement examples it will be appreciated by those skilledin the art that the amplifier 102 may in whole or in part be included aspart of the headphones.

FIG. 23 b illustrates a variation of part of the amplifier 102illustrated in FIG. 23 a.

FIG. 23 b illustrates an amplifier 102 that receives two input controlsignals S21′ and S22′ that respectively control gain controllers (notillustrated) within the gain unit 220. The two input control signalsS21′ and S22′ are also fed into a controller unit 230 that detects themaximum value of the two control signals S21′ and S22′ such that thevariable voltage power supply 80, and therefore the supply voltages+/−Vout, is varied in response to the greater of the two volume controlsignals S21′ and S22′. The two control signals may represent separatevolume control signals in an application where either a balance controlor separate volume controls for each input signal is required.

An alternative arrangement (not illustrated) to that of FIG. 23 b iswhere the control signals S21′ and S22′ each control a variable voltagepower supply that supplies power to the respective output stage to whichthe control signal relates.

A further alternative arrangement (not illustrated) to that of FIG. 23 bis where the gain unit 220 or respective elements thereof are fully orpartially incorporated into the output stage unit 405 or respectiveelements thereof.

FIG. 24 illustrates the amplifier 100 with different transducers, i.e.loads, that represent non-exhaustive illustrations of basic applicationsfor the novel amplifier. It will be appreciated that illustrated exampleembodiments of this FIG. 24 , and FIG. 25 , are equally applicable todual ground referenced voltage systems such as illustrated in FIG. 2 a.

In a first of two illustrated examples of FIG. 24 , the amplifier 100may be employed in an audio system, such as: a portable music system(MP3) (including such devices combined with mobile telephone handsets orsimilar devices); Hi-Fi; In-Car Entertainment system; or a DVD playerfor example, whereby the system receives a volume, or level, controlsignal S2′ that is altered by a user either via a potentiometer, i.e. avolume knob, or by a remote control device for example. In thisparticular example of an application, the output signal S5 of theamplifier 100 is used to drive a speaker SP. It will be appreciated tothose skilled in the art that in modern audio systems it is quite usualto have a plurality of output signals such as, for example, in stereosystems or Dolby® pro logic 5.1 channel surround sound systems.

In a second illustrated example, the amplifier 100 may be employed in atransmitter system such as a mobile phone RF transmitter, whereby itreceives a transmit power control signal S2′. In this particular exampleof an application, the output signal S5 of the amplifier 100 is used todrive a transmitter TR, such as an aerial for example.

FIG. 25 illustrates a data transmitter/receiver system, such as a modemfor example, wherein an amplifier 102 acts as a line driver for the datatransmission/receiver system. A power supply 200, gain controller 20,line driver 401, system controller 700, signal modulator 710, signaldemodulator 720, a transmit/receive controller 730 and transmission line740 are arranged as shown. Again the amplifier 102 works in the same wayas the previous examples with the variable voltage power supplysupplying the line driver 401 with a dynamic voltage +/−Vout. Thecontrol signal S2′ from the system controller 700 controls the gaincontroller 20 and the voltage level of the dynamic voltage +/−Vout. Themodulator 710, which is also controlled by the system controller 700,provides the input signal S1 to the amplifier. The transmit/receivecontroller 730 allows for two-way signal transmission between the outputof the amplifier 102 and the transmission line 740. Controller 730 maybe a two-to-four wire hybrid to allow full duplex, or a switchingelement to allow transmission in one direction at a time. Thetransmit/receive controller 730 also allows a received signal to be fedback to the data transmitter/receiver system via the demodulator 720.

Such a novel amplifier as herein described may be implemented usingdiscrete components or may be implemented on an integrated circuit or acombination of both.

It should be noted that the above described embodiments illustraterather than limit the invention, and that those skilled in the art willbe able to design many alternative embodiments without departing fromthe scope of the appended claims and drawings. The word “comprising”does not exclude the presence of elements or steps other than thoselisted in a claim, “a” or “an” does not exclude a plurality (unlesscontext requires otherwise), and a single element may fulfil thefunctions of several elements recited in the claims. It should also benoted that the attenuation, or decrease, of a signal amplitude is a formof amplification, thus the word “amplify”, amplifying”, “amplified” andthe like can be taken to mean an increase or a decrease in the amplitudeof a signal. Any reference signs in the claims shall not be construed soas to limit their scope.

What is claimed is:
 1. A wireless headphone comprising: a wirelessreceiver for receiving a wireless signal comprising audio data; a firstamplifier configured to receive a first input audio signal from thewireless receiver and output a first output audio signal to drive afirst speaker; and a charge pump configured to provide a supply voltageto the first amplifier, wherein the charge pump is operable so that amagnitude of the supply voltage is variable based on an indication of anamplitude of the first input audio signal.
 2. The wireless headphone asclaimed in claim 1 wherein said wireless signal comprises an RF signal.3. The wireless headphone as claimed in claim 1 further comprising adigital signal processor for processing audio data received from thewireless receiver to provide said first input audio signal.
 4. Thewireless headphone as claimed in claim 1 further comprising a digital toanalogue converter configured to receive a first digital audio inputsignal and to output an analogue audio signal as said first input audiosignal.
 5. The wireless headphone as claimed in claim 1 furthercomprising a second amplifier configured to receive a second input audiosignal from the wireless receiver and output a second output audiosignal to drive a second speaker.
 6. The wireless headphone as claimedin claim 5 wherein the charge pump is further configured to provide saidsupply voltage to the second amplifier and wherein the charge pump isoperable so that the magnitude of the supply voltage is further variablebased on an indication of an amplitude of the second input audio signal.7. The wireless headphone as claimed in claim 1 wherein the charge pumpis configured to receive an input voltage and is operable to generatepositive and negative supply voltages with respect to a common voltage.8. The wireless headphone as claimed in claim 7 wherein the charge pumpis selectively operable in: a first mode in which the positive andnegative supply voltages each have a magnitude equal to a magnitude ofthe input voltage; and a second mode in which the positive and negativesupply voltages each have a magnitude equal to half the magnitude of theinput voltage.
 9. The wireless headphone as claimed in claim 8 whereinthe charge pump comprises a positive supply output node for outputtingsaid positive supply voltage; a negative supply output node foroutputting said negative supply voltage; a plurality of flying capacitornodes for connecting to at least one flying capacitor; a switch networkfor selectively connecting said positive supply output node; saidnegative supply output node and said flying capacitor nodes in asequence of states; and a switch controller for operating said switchnetwork in a sequence of states, in use with a single flying capacitorconnected to said flying capacitor nodes, to selectively provide saidfirst mode or said second mode.
 10. The wireless headphone as claimed inclaim 1 further comprising a gain controller for controlling a gainapplied to the first audio first input audio signal or the first outputaudio signal.
 11. The wireless headphone as claimed in claim 10 whereinthe charge pump is operable so that the magnitude of the supply voltageis variable based on a value of gain applied by the gain controller. 12.The wireless headphone as claimed in claim 1 wherein the wirelessheadphone comprises an earbud.
 13. A wireless headphone set comprisingfirst and second headphones, wherein each of the first and secondheadphones is a wireless headphone as claimed in claim
 1. 14. Thewireless headphone set of claim 13 wherein each of the first and secondheadphones comprises an earbud.
 15. A wireless headset comprising: afirst amplifier configured to amplify a first audio signal to generate afirst audio driving signal to drive a first audio output transducer; anda charge pump configured to provide a supply voltage to the firstamplifier, wherein the charge pump is operable so that a magnitude ofthe supply voltage is variable; wherein the first audio signal isderived a wireless signal received wirelessly by the wireless headset.16. The wireless headset as claimed in claim 15 further comprising avoltage controller for controlling the magnitude of said supply voltagebased on an indication of amplitude of the first audio driving signal.17. The wireless headset as claimed in claim 15 further comprising asecond amplifier configured to amplify a second audio signal to generatea second audio driving signal to drive a second speaker; wherein thesecond audio signal is also derived a wireless signal receivedwirelessly by the wireless headset.
 18. The wireless headset as claimedin claim 17 wherein the charge pump is further configured to providesaid supply voltage to the second amplifier.
 19. The wireless headset asclaimed in claim 15 wherein the charge pump is selectively operable in:a first mode to generate positive and negative supply voltages eachhaving a magnitude equal to a magnitude of an input voltage; and asecond mode to generate positive and negative supply voltages each havea magnitude equal to half the magnitude of said input voltage.
 20. Awireless earbud comprising: a receiver for receiving a wireless signalto generate a first audio signal; an amplifier configured to receive thefirst audio signal and generate a first driving signal to drive a firstaudio transducer; and a variable voltage power supply for supplying avariable supply voltage to the output driver stage based on anindication of an amplitude of the first audio signal.